The SAW convolver is widely utilized as a central element in the receiving section of an SS communication apparatus. FIGS. 7A and 7B show block diagrams indicating the fundamental operation (by a prior art method) of the SS communication apparatus.
FIG. 7A represents a transmitter, in which reference numeral 1 is a data signal; 2 is a PN code generator; 3 is a mixer; 4 is a carrier signal; 5 is an amplifier; and 13 is 6 is a transmitting antenna. FIG. 7B represents a receiver, in which 7 is a receiving antenna; 8 is a local oscillator; 9 is an oscillator; 10 is an SAW convolver; 11 is a low pass filter; 12 is a high pass filter; and a data demodulator.
By the prior art method indicated in FIGS. 7A and 7B, data which are to be transmitted are mixed with a PN code to be transmitted. Since the frequency band of the PN code is wider than the frequency band of the data signal, the spectrum of the transmitted signal E(t) is spread over a frequency band wider than the spectrum of the original data signal D(t). On the other hand, the receiving section indicated in FIG. 7B is so constructed that the received signal S(t) and an internal reference signal R(t) are subjected to a convolution integration by means of the SAW convolver 10. The signal S(t) is a signal obtained by converting the frequency of the received signal by means of the local oscillator 8 and the mixer 3 so that the frequency of the received signal is contained in the working frequency band (inputted frequency band) of the SAW convolver. On the other hand, the reference signal R(t) is a signal obtained by spreading the sinusoidal wave having a frequency f.sub.r determined similarly within the inputted frequency band of the SAW convolver. f.sub.c representing the carrier frequency of the transmitted signal, the frequency f.sub.L of the local oscillator 8 is set usually so that f.sub.L =f.sub.c -f.sub.r. Further the PN code within the receiver is set so as to be a code, which is the mirror image (code B) of a certain PN code (code B). In the case where the codes are set in such a way, a code A representing the PN code of the transmitter, the magnitude of the output of the SAW convolver 10 in the receiver .vertline.C(t).vertline. is proportional to the magnitude of the correlation signal of the code A and the code B. That is, the SAW convolver 10 is used as an element executing the correlation operation between the PN codes set by the transmitter and the receiver. If the PN codes set by the transmitter and the receiver are identical to each other (B=A), a strong correlation output appears in the output of the SAW convolver and thus the communication between the transmitter and the receiver is possible. In the case where the PN codes are different from each other or a signal of the prior art narrow frequency band communication is inputted, only a small correlation output appears in the SAW convolver and thus the communication is impossible.
Consequently it is possible to select a communication party by selecting a PN code and in addition to realize a communication having an extremely small mutual interference with the prior art narrow frequency band communication. In the case where the PN codes are in accordance with each other and thus the communication is possible, information of the data signal 1 appears in the form of variations in the phase or the amplitude of the convolver output C(t). The data demodulator 13 is a section, in which the transmitted data are restored, starting from the phase or the amplitude of the convolver output C(t). The concrete construction of the data demodulator 13 differs, depending on the type of modulation (FSK, DPSK, etc.) of the data signal and further it differs also, depending on what kind of convolver is used, a usual one-track convolver, a 2-gate SAW convolver, or a 2-track SAW convolver. However, as described later, the object of the present invention is to improve the correlation processing method by an SAW convolver and it doesn't relate to the data demodulation method. Further FIGS. 7A and 7B indicate the basic operation of an SS communication apparatus using an prior art SAW convolyer in a simple manner, in which an AGC, a code synchronizing section, etc. are not indicated. The content of the prior art method is described more in detail in the following literatures.
Literature [1]:
Tsubouchi, et al.; "Asynchronous type SSC transceiver using an SAW convolver" Spread Spectrum Communication Study Group of Electronic Information Communication Society of Japan; Technical Report of Electronic Information Communication Society, Vol. 2, No. 1, SS 88-7, April, 1988, pp 40.about.47.
Literature [2]:
Hamatsu, et al.; "Packet type spread spectrum wireless MODEM using an SAW convolver"; Spread Spectrum Communication Study Group of Electronic Information Communication Society of Japan; Technical Report of Electronic Information Communication Society, Vol. 2, No. 1, SS 88-8, April, 1988, pp 48.about.53.
Literature [3]:
Mori, et al.; "Code synchronization holding method in a spread spectrum receiver using an SAW convolver"; Report Collection of Electronic Communication Society of Japan, Vol. J69-B, No. 4, 1986, pp 404.about.405.
An SS communication apparatus using an SAW convolver as described above has various advantages with respect to other SS communication apparatuses. For example, with respect to the SS communication method using a sliding correlator, it has an advantage that the initial synchronization catching time is extremely short. Further, with respect to the SS communication method using a matched filter, it has an advantage that the PN codes can be switched in real time.
However the SS communication apparatus using an SAW convolver by the prior art method has various problems caused by the fact that the SAW convolver is no ideal convolution integrator. A problem, which is particularly serious among them, is an influence of the self convolution signal produced in the SAW convolver. FIG. 8 is a top view of the SAW convolver for explaining the self convolution in a simple manner, in which reference numeral 14 is a piezo-electric substrate; 15 is an interdigital electrode; 16 is a gate electrode; 17 is an input terminal; and 18 an output terminal.
In FIG. 8, when electric signals S(t) and R(t) are applied to the two input terminals 17, they are converted into SAWs at the respective interdigital electrodes 15. At this time, if only surface waves .phi..sub.g and .phi..sub.r, which propagate in two directions opposite to each other, existed on the gate electrode, the signal appearing on the output terminal would represent an ideal convolution integral (excepting that the integration time is finite). However, in reality, the surface waves .phi..sub.g and .phi..sub.r are reflected by the interdigital electrodes 15, which are opposite to each other, which gives rise to reflected waves .phi..sub.g ' and .phi..sub.r '. In such a case the output signal can be expressed by the following formula; ##EQU1## where L represents the gate length; v denotes the propagation speed of the SAW; K is a constant; .GAMMA. is a reflection coefficient of the SAW by the interdigital electrodes; and .alpha. is a attenuation constant of the SAW, and further T corresponds to the in-gate delay time.
The first term of Eq. (1) represents the convolution integration signal of S(t) and R(t) and as indicated by Eq. (1), also signals expressed by the second and third terms, other than the first term, take place. The second and third terms are terms, which would not exist, if there were no reflected wave (.GAMMA.=0), and represent a self convolution signal. It will be clear that, when there exist such influences of the self convolution signal, the correlation output C(t) in the receiver indicated in FIG. 7B is influenced thereby. One of these influences is that the spurious level of the correlation output becomes higher than the ideal value and that as the result the rate of errors at the restoration of the data increases. Another of the influences is that, in the case where a strong narrow frequency band signal is mixed in the input signal S(t) through the receiving antenna 7, the spurious level of the correlation output is raised similarly and that the rate of errors at the restoration of the data increases remarkably. FIGS. 9A to 9E show this aspect.
FIG. 9A indicates the input signal, in the case where there are no disturbing signals N(t); FIG. 9B the reference signal; FIG. 9C a narrow frequency band disturbing signal; FIG. 9D the correlation output, in the case where there are no disturbing signals N(t); and FIG. 9E the correlation output, in the case where disturbing signals N(t) are mixed.
FIGS. 9A to 9E show an example indicating how the correlation output C(t) varies, in the case where a strong narrow frequency band signal N(t) is mixed in the input signal S(t). Further FIGS. 9A to 9E show an example, in the case where the period of the PN code is equal to the in-gate delay time T.
In the case where a narrow frequency band disturbing signal N(t) having a frequency f.sub.1 is mixed in the input signal, a self convolution output of N(t) having a frequency 2f.sub.i appears in the correlation output C(t), as indicated in FIG. 9E, and as the result, the effective spurious level is raised. Further the correlation signal and the self convolution signal of N(t) interfere with each other, which gives rise to an effect that the correlation peak is AM modulated by a frequency 2.vertline.f.sub.r -f.sub.i .vertline.. These two effects leads to a result that the rate of errors at the restoration of the data is increased remarkably.
That is, the prior art method indicated in FIGS. 7A and 7B has drawbacks that influences of the self convolution signal generated in the SAW convolver are inevitable and that the rate of errors at the restoration of the data increases, exceeding the ideal value. In particular, when a strong narrow frequency band signal is mixed in the input signal, the rate of errors is increased and this point is one of the most serious drawbacks in practice.